Not applicable.
Not applicable.
The field of the invention is AC induction motor drives and more specifically the area of injecting high frequency voltage signals into an AC induction motor and using high frequency feedback signals to identify dynamic motor operating parameters.
Induction motors have broad application in industry, particularly when large horsepower is needed. In a three-phase induction motor, three phase alternating voltages are impressed across three separate motor stator windings and cause three phase currents therein. Because of inductances, the three currents typically lag the voltages by some phase angle. The three currents produce a rotating magnetic stator field. A rotor contained within the stator field experiences an induced current (hence the term xe2x80x9cinductionxe2x80x9d) which generates a rotor field. The rotor field typically lags the stator field by some phase angle. The rotor field is attracted to the rotating stator field and the interaction between the two fields causes the rotor to rotate.
A common rotor design includes a xe2x80x9csquirrel cage windingxe2x80x9d in which axial conductive bars are connected at either end by shorting rings to form a generally cylindrical structure. The flux of the stator field cutting across the conductive bars induces cyclic current flows through the bars and across the shorting rings. The cyclic current flows in turn produce the rotor field. The use of this induced current to generate the rotor field eliminates the need for slip rings or brushes to provide power to the rotor, making the design relatively maintenance free.
To a first approximation, the torque and speed of an induction motor may be controlled by changing the frequency of the driving voltage and thus the angular rate of the rotating stator field. Generally, for a given torque, increasing the stator field rate will increase the speed of the rotor (which follows the stator field). Alternatively, for a given rotor speed, increasing the frequency of the stator field will increase the torque by increasing the slip, that is the difference in speed between the rotor and the stator fields. An increase in slip increases the rate at which flux lines are cut by the rotor, increasing the rotor generated field and thus the force or torque between the rotor and stator fields.
Referring to FIG. 1, a rotating phasor 1 corresponding to a stator magneto motive force (xe2x80x9cmmfxe2x80x9d) will generally have some angle xcex1 with respect to the phasor of rotor flux 2. The torque generated by the motor will be proportional to the magnitudes of these phasors 1 and 2 but also will be a function of their angle xcex1. Maximum torque is produced when phasors 1 and 2 are at right angles to each other whereas zero torque is produced if the phasors are aligned. The stator mmf phasor 1 may therefore be usefully decomposed into a torque producing component 3 perpendicular to rotor flux phasor 2 and a flux component 4 parallel to rotor flux phasor 2.
These two components 3 and 4 of the stator mmf are proportional, respectively, to two stator current components: iq, a torque producing current, and id, a flux producing current, which may be represented by orthogonal vectors in the rotating frame of reference (synchronous frame of reference) of the stator flux having slowly varying magnitudes.
Accordingly, in controlling an induction motor, it is generally desired to control not only the frequency of the applied voltage (hence the speed of the rotation of the stator flux phasor 1), but also the phase of the applied voltage relative to the current flow and hence the division of the currents through the stator windings into the iq and id components. Control strategies that attempt to independently control currents iq and id are generally termed field oriented control strategies (xe2x80x9cFOCxe2x80x9d).
Generally, it is desirable to design FOCs that are capable of driving motors of many different designs and varying sizes. Such versatility cuts down on research, development, and manufacturing costs and also results in easily serviceable controllers.
Unfortunately, while versatile controllers are cost-effective, FOC controllers cannot control motor operation precisely unless they can adjust the division of d and q-axis currents through the stator windings to account for motor-specific operating parameters. For this reason, in order to increase motor operating precision, various feedback loops are typically employed to monitor stator winding currents and voltages and/or motor speed. A controller uses feedback information to determine how the inverter supplied voltage must be altered to compensate for system disturbances due to system specific and often dynamic operating parameters and then adjusts control signals to supply the desired inverter voltages.
To this end, in an exemplary FOC system two phase d and q-axis command currents are provided that are calculated to control a motor in a desired fashion. The command currents are compared to d and q-axis motor feedback currents to generate error signals (i.e., the differences between the command and feedback currents). The error signals are then used to generate d and q-axis command voltage signals which are in turn transformed into three phase command voltage signals, one voltage signal for each of the three motor phases. The command voltage signals are used to drive a pulse width modulated (PWM) inverter that generates voltages on three motor supply lines. To provide the d and q-axis current feedback signals the system typically includes current sensors to sense the three phase line currents and a coordinate transformation block is used to transform the three phase currents to two phase dq frame of reference feedback currents.
In addition to requiring two phase signals and three phase signals to perform 2-to-3 and 3-to-2 phase transformations, respectively, a flux position angle is also required. One common way to generate a flux angle feedback estimate is to integrate a stator frequency. A stator frequency can be determined by adding a measured rotor frequency (rotor speed) and calculated slip frequency. Slip frequency calculations require motor parameter values such as a rotor resistance, leakage inductance, etc. Therefore, precise parameter values are necessary to eliminate errors in flux angle determinations.
In the case of drives that do not include a speed sensor it is necessary to estimate both the rotor frequency and the slip frequency to determine the flux angle. Thus, these drives also require precise knowledge of motor parameter values. Usually, drives that do not includes speed sensors cannot operate at low rotor frequencies (e.g., below 3 Hz) due to motor parameter estimation errors.
Recently, a method for flux angle feedback determination was developed to overcome the low speed operating problems in drives that do not include speed sensors. This method includes injecting a known high frequency voltage signal into each of the command voltage signals used to drive the PWM inverter and use feedback current (or voltage) signals to determine the flux angle. To this end, an exemplary system converts a high frequency command signal into a high frequency phase angle and then generates a first injection signal that is the product of a scalar and the sine of the high frequency phase angle. Second and third injection signals are also generated, each of the second and third signals phase shifted from the first signal by 120 degrees. A separate one of the first, second and third signals is then added to a separate one of the three voltage command signals.
Algorithms to generate a flux position angle estimate as a function of a negative sequence high frequency current components or zero sequence high frequency current (or voltage) components are well known in the controls art and therefore will not be explained here in detail.
The high frequency voltage injection methods can be used to identify a flux angle position without using motor parameters. The derivative of the flux angle position can be used as an estimated stator frequency. Therefore, an estimated rotor speed can be determined by subtracting slip frequency from estimated stator frequency. As well known, slip frequency is a function of motor parameters. Thus, to accurately estimate speed it is still necessary to precisely identify motor parameters such as rotor resistor, etc. In many cases nameplate resistances and inductances are provided for specific motors and those values can be used in system algorithms to generate the estimates.
Unfortunately, several of the parameters that are used in the speed estimation algorithms change as a function of motor operation and therefore the algorithms are not always accurate and, in fact, the accuracy of the algorithms changes during motor operation. For instance, stator resistance changes with temperature and stator winding temperature increases as the average current through a winding increases. Thus, in a variable speed motor, when inverter supplied voltage and current is increased or decreased, the stator resistance is subject to change which in turn results in an unexpected demand in stator voltage. Similar comments are applicable to rotor resistance.
Recognizing these problems the industry has devised various methods of determining stator and rotor resistances during motor operation. Unfortunately, each of the prior methods have been either extremely computationally intensive, relatively inaccurate or relatively expensive to implement (e.g., requiring a plurality of sensors, etc.). Therefore, it would be advantageous to have a relatively inexpensive and simple method and apparatus that could generate accurate stator and rotor resistance estimates for use in induction motor control with or without speed sensor.
The angle between a high frequency injection voltage and a resulting high frequency current can be used to determine an instantaneous stator resistance value. The stator resistance value along with some commissioning values can in turn be used to determine a rotor resistance value. Thus, if the voltage-current lag angle between high frequency voltage and current can be determined dynamically and routinely, the rotor and stator resistances can be determined.
It has been recognized that using a voltage-current lag angle estimate, three phase injected voltages can be mapped to a dq high frequency voltage reference frame wherein the high frequency voltage has to be aligned with a d axis and three phase feedback currents can be mapped to a new established current dq reference frame wherein the high frequency current is aligned with one of the d or q axis when the estimate is accurate. Inaccuracy in the estimate shows up as a vector component orthogonal to the vector with which the high frequency current should be aligned. Therefore, by adjusting the voltage-current lag angle estimate as a function of the orthogonal vector component, the voltage-current lag angle estimate error can be eliminated. Specifically, by driving the orthogonal vector component to zero, an accurate angle estimate can be rendered. Thereafter the angle estimate can be used to determine instantaneous stator and rotor resistance estimates.
An exemplary embodiment of the invention includes a method for estimating a motor stator resistance in a three phase motor characterized by a stator leakage inductance and driven by a pulse width modulator that is linked to the motor via three supply lines, the method for use with a system that receives a high frequency voltage command signal and causes high frequency voltages to be injected into each of the three motor phases at the high frequency. Let""s assume for simplicity that the high frequency voltage is injected along the d-axis. Here, the method comprises the steps of identifying a high frequency command angle, subtracting a current lag angle estimate from the high frequency command angle to generate a high frequency current angle, obtaining feedback currents from the supply lines, converting the feedback currents into two orthogonal phase currents using the high frequency current angle, providing a closed loop system for a first of the two orthogonal phase currents to drive the first of the two orthogonal phase currents to zero by using a PI controller, using the output of this PI controller as a current lag angle estimate and using the current lag angle estimate as a voltage-current lag angle estimate to generate the stator resistance estimate.
The step of using the voltage-current lag angle estimate may include the step of mathematically combining the angle estimate, the stator leakage inductance and the high frequency command signal to generate the stator resistance estimate. The step of mathematically combining in some embodiments includes the step of determining the cotangent of the voltage-current lag angle estimate to generate a cotangent value, converting the high frequency command signal into a radians per second value and multiplying the radians per second value, the cotangent value and the stator leakage reactance to generate the stator resistance estimate.
The invention is also for generating a rotor resistance estimate and, to that end, may include the step of using the voltage-current lag angle estimate to generate the rotor resistance estimate. In this case, the step of using the voltage-current lag angle estimate to generate the rotor resistance estimate in some embodiments includes the step of identifying and storing an initial voltage-current lag angle estimate, identifying an initial rotor resistance estimate, dividing the initial voltage-current lag angle estimate by the voltage-current lag angle estimate to generate a stator resistance ratio and multiplying the initial rotor resistance estimate by the stator resistance ratio to generate the rotor resistance estimate.
The step of mathematically combining may include the steps of determining the tangent of ninety degrees less the voltage-current lag angle estimate to generate a tangent value, multiplying the high frequency command signal in radians per second, the tangent value and the stator leakage reactance to generate the stator resistance estimate.
The step of identifying a voltage lag angle estimate where the voltage initially does not align with the d axis may include the steps of converting the high frequency signal into a high frequency command angle, subtracting a voltage lag angle estimate from the high frequency command angle to generate a high frequency voltage angle, obtaining feedback voltages from the supply lines, converting the feedback voltages into two orthogonal phase voltages using the high frequency voltage angle, providing a closed loop system for a first of the two orthogonal phase voltages to drive the first of the two orthogonal phase voltages to zero by using of a PI controller, using the output of this PI controller as a voltage lag angle estimate and then use this voltage lag angle estimate for the voltage-current lag angle estimate calculation by subtracting the voltage lag angle estimate from the current lag angle estimate.
The invention also includes apparatus for performing each of the methods described above.
Thus, it should be appreciated that relatively inexpensive and simple systems and methods have been described for determining the phase lag angle between high frequency voltage and current signals which can be performed essentially continuously during motor operation or on a periodic basis, the lag estimate then useable to determine rotor and stator resistances to facilitate more accurate motor control.
These and other objects, advantages and aspects of the invention will become apparent from the following description. In the description, reference is made to the accompanying drawings which form a part hereof, and in which there is shown a preferred embodiment of the invention. Such embodiment does not necessarily represent the full scope of the invention and reference is made therefore, to the claims herein for interpreting the scope of the invention.